Methods and apparatus for reducing the average-to-minimum magnitude ratio of communications signals in communications transmitters

ABSTRACT

A communications transmitter configured to reduce the average-to-minimum magnitude ratio (AMR) of a communications signal includes a symbol mapper, a pulse-shaping filter, an AMR reduction circuit, and a modulator. The symbol mapper operates to generate a sequence of symbols from a binary-source data stream containing a message to be transmitted, and the pulse-shaping filter generates a baseband signal based on the sequence of symbols. The AMR reduction circuit is configured to compare magnitudes of samples of the baseband signal to a time-varying low-magnitude threshold, and modify the baseband signal to produce a modified baseband signal having a reduced AMR if a sample of the baseband signal is determined to have a magnitude less than the time-varying low-magnitude threshold. Finally, the modulator operates to modulate a carrier signal based on the modulation information contained in the modified baseband signal.

FIELD OF THE INVENTION

The present invention relates to communications systems and methods.More specifically, the present invention relates to methods andapparatus for reducing the average-to-minimum magnitude ratio (AMR) ofcommunications signals in communications transmitters.

BACKGROUND OF THE INVENTION

A key and essential component of every radio frequency (RF)communications system is the RF transmitter. As shown in FIG. 1, an RFtransmitter 100 generally comprises a baseband modulator 102, afrequency upconverter 104, a power amplifier (PA) 106, and an antenna108. The purpose of the baseband modulator 102 is to generate a basebandsignal s(t) that contains a message to be transmitted and which isformatted in accordance with a predetermined modulation scheme. Thepurpose of the frequency upconverter 104 is to upconvert the basebandsignal s(t) to RF, so that the message is capable of being propagatedthrough space (i.e., transmitted over the air) to a remote receiver. ThePA 106 is employed to increase the power of the RF signal before it isradiated by the antenna 108, thereby compensating for attenuation of theRF signal as it is transmitted over the air to the remote receiver.

In modern RF transmitters, the message to be transmitted is included ina binary-source data stream. The baseband modulator 102 groups the databits in the binary-source data stream into a sequence of N-bit words,where N is some positive integer, and maps the pattern of bits in eachN-bit word to one of M=2^(N) possible symbols. The M symbols define themodulation scheme being employed, and affect how the amplitude and/orangle of the RF carrier signal is varied (i.e., modulated) to carry themessage in the original binary-source data stream to the remotereceiver. By mapping each N-bit word to one of M possible symbols,N=log₂M bits can be transmitted in each symbol.

The symbols generated by the baseband modulator 102 comprise a sequenceof weighted impulses. To limit the bandwidths of the impulses, thebaseband modulator 102 is further configured to shape each impulse by aband-limiting pulse p(t).

Mathematically, the baseband signal s(t) can be expressed as:

${{s(t)} = {\sum\limits_{n}\;{a_{n}{p\left( {t - {nT}_{s}} \right)}}}},$where n is a symbol index, a_(n) is the n^(th) symbol in the sequence ofsymbols, p(t) is the pulse at time t, and T_(s) is the symbol period.Each a_(n) is either a real or complex number having one of M possiblestates. For example, in the quadrature phase-shift keying (QPSK)modulation scheme, M=4, and a_(n) is given by a_(n)=e^(jπ(2d) ^(n)^(+1)/2), where d_(n) is an integer selected from the set {0, 1, 2, 3}.

Because the baseband signal s(t) is in general a complex signal, it isusually expressed in terms of its in-phase (I) and quadrature (Q)components, i.e., as s(t)=I(t)+jQ(t), and the baseband modulator 102 isconfigured to generate separate pulse-shaped I and Q baseband signalsfor each of the I and Q channels. This is illustrated in FIG. 2, whichis simplified drawing of a typical RF transmitter 200. The RFtransmitter 200 comprises a baseband modulator 202; I-channel andQ-channel digital-to-analog converters (DACs) 204 and 206; a transmitlocal oscillator (Tx-LO) 208; a quadrature modulator 210; a PA 212; andan antenna 214. Due to its use of the quadrature modulator 210, the RFtransmitter 200 is referred to in the description below as the“quadrature-modulator-based” transmitter 200.

The quadrature modulator 210 of the quadrature-modulator-basedtransmitter 200 includes an I-channel mixer 216, a Q-channel mixer 218,a ninety-degree phase shifter 220, and a combiner 222. The I-channel andQ-channel DACs 204 and 206 convert the pulse-shaped I and Q basebandsignals from the baseband modulator 202 into analog I and Q basebandsignals. The quadrature modulator 210 then upconverts the analog I and Qbaseband signals to RF. Specifically, the I-channel mixer 216 mixes theanalog I baseband signal with an RF carrier signal provided by the Tx-LO208, while the Q-channel mixer 218 mixes the analog Q baseband signalwith a ninety-degree phase-shifted version of the RF carrier signalproduced at the output of the ninety-degree phase shifter 220, therebyproducing upconverted I- and Q-channel RF carrier signals. Theupconverted I- and Q-channel RF carrier signals are then combined by thecombiner 222 to produce the desired modulated RF carrier signal, whichis finally amplified by the PA 212 and radiated over the air to a remotereceiver by the antenna 214.

The quadrature-modulator-based RF transmitter 200 is satisfactory formany applications. However, it is not the most desirable solution whenused in communication systems employing non-constant envelope modulationschemes. Current and next generation mobile telecommunications systemscommonly employ non-constant envelope modulation schemes to achievehigher data rates for a given bandwidth of the RF spectrum than can berealized using constant envelope modulation schemes. However, asexplained below, their use in quadrature-modulator-based RF transmittersrequires a sacrifice of energy efficiency.

A non-constant envelope modulation scheme produces a baseband signalthat has a non-constant (i.e., time-varying) envelope. Consequently, andas illustrated in FIG. 3, the modulated RF carrier signal presented tothe RF input RFin of the PA 212 also has a non-constant envelope. Toprevent the PA 212 from clipping the signal peaks of the modulated RFcarrier signal, the input signal power to the PA 212 must be reduced.This technique, known as power back-off, ensures that the PA 212 alwaysoperates in its linear region of operation, even during times when themagnitude of the modulated RF carrier signal is at its peak.Unfortunately, while power back-off helps to ensure linearity it alsoundesirably results in a reduction in energy efficiency.

The energy efficiency of an RF transmitter is determined in large partby how efficient the RF transmitter's PA is, since the PA is usually thedominant consumer of energy in the RF transmitter. The energy efficiencyof the PA is determined by the ratio of the PA RF output power to thedirect current (DC) power supplied to the PA from the RF transmitter'spower supply. Consequently, when power back-off is employed the energyefficiency of the RF transmitter is reduced. The reduction in energyefficiency is most severe for signals that have a high peak-to-averageratio (PAR). Unfortunately, many modern non-constant envelope schemesproduce signals having high PARs.

An RF transmitter having low energy efficiency is undesirable in mostany circumstance. It is particularly undesirable when the RF transmittercomprises a battery-powered RF transmitter, such as used in a cellularhandset, since the low energy efficiency results in shortened batterylife. Fortunately, an alternative type of communications transmitterknown as a polar transmitter is available which avoids the linearityversus energy efficiency tradeoff that plagues thequadrature-modulator-based transmitter 200. In a polar transmitter theamplitude information (i.e., the signal envelope) is temporarily removedfrom the non-constant envelope signal while the remaining signal, whichhas a constant envelope, is upconverted to RF. As explained in moredetail below, the previously removed signal envelope is used to modulatethe power supplied to the PA as the upconversion process takes place.Because the signal applied to the RF input of the PA has a constantenvelope, a more energy efficient nonlinear PA can be used without therisk of signal peak clipping.

FIG. 4 is a drawing showing the salient elements of a typical polartransmitter 400. The polar transmitter 400 comprises a basebandmodulator 402; a Coordinate Rotation Digital Computer (CORDIC) converter(i.e., rectangular-to-polar converter) 404; an amplitude path includingan amplitude path DAC 406 and amplitude modulator 408; an angle pathincluding an angle path DAC 410 and angle modulator 412; a PA 414; andan antenna 416. The purpose of the CORDIC converter 404 is to convertthe digital rectangular-coordinate pulse-shaped I and Q baseband signalsfrom the baseband modulator 402 to digital polar-coordinate amplitudeand angle component signals p and θ. The amplitude and angle path DACs406 and 410 convert the digital amplitude and angle component signals ρand θ into analog amplitude and angle modulation signals. In theamplitude path, the amplitude modulator 408 then modulates a directcurrent power supply voltage Vsupply (e.g., as provided by a battery) bythe amplitude information in the analog amplitude modulation signal. Theresulting amplitude-modulated power supply signal Vs(t) is supplied tothe power supply port of the PA 414. Meanwhile, in the angle path, theangle modulator 412 operates to modulate an RF carrier signal by theangle information in the analog angle modulation signal, therebyproducing an angle-modulated RF carrier signal which is coupled to theRF input port RFin of the PA 414.

As shown in FIG. 5, the angle-modulated RF carrier signal at the RFinput port RFin of the PA 414 has a constant envelope. This permits thePA 414 to be configured to operate in its nonlinear region of operation(i.e., as a “nonlinear” PA) without the risk of signal peak clipping, aswas mentioned above, and the PA 414 can be operated without having toback-off the output power. Typically the PA 414 is implemented as ahighly-efficient switch-mode PA (e.g., as a Class D, E or F switch-modePA) switching between compressed and cut-off states. When configured inthis manner, the amplitude-modulated power supply signal Vs(t) modulatesthe power supply port of the PA 414 and the envelope information isrestored to the RF output RFout of the PA 414 as the PA 414 amplifiesthe angle-modulated RF carrier signal. By operating the PA 414 as aswitch and dynamically controlling the power supplied to it, the polartransmitter 400 is able to achieve significantly higher energyefficiencies than the quadrature-modulator-based RF transmitter 200.

Although the polar transmitter 400 is able to handle non-constantenvelope signals at higher energy efficiencies than the moreconventional quadrature-modulator-based transmitter 200, the amplitudeand angle component signals ρ and θ typically have much higherbandwidths compared to the rectangular-coordinate I and Q basebandsignals from which they derive. This so-called “bandwidth expansion”phenomenon occurs during the rectangular-to-polar conversion processperformed by the CORDIC converter 404. The high bandwidths aremanifested as high-frequency events in the amplitude and angle componentsignals ρ and θ and are highly undesirable. Not only do thehigh-frequency events tend to degrade the modulation accuracy of thepolar transmitter 400, they also cause the transmission spectrum toextend beyond its intended band-limited channel, resulting in adjacentchannel interferers and an increase in receive band noise. These effectscan be very difficult to deal with, especially when strict noiserestriction standards must be adhered to.

The extent to which high-frequency events end up appearing in theamplitude and angle component signals ρ and θ is very much dependent onthe modulation scheme that is employed. In particular, non-constantmodulation schemes that produce signals having a high average-to-minimummagnitude ratio (AMR) generally have a very large angle componentbandwidth. In fact, for modulation schemes that produce signals whichpass through zero, as illustrated in the signal trajectory diagram inFIG. 6, phase changes by as much as 180 degrees can occur, resulting inan angle component signal θ having essentially infinite bandwidth.Signals of such high bandwidth cannot be accurately processed andtransmitted by the polar transmitter 400, or by any type of transmitterfor that matter, and the high-frequency content in such signals makesstandards compliance extremely difficult, and in some cases impossible,to achieve.

Various techniques have been proposed to reduce high-frequency events inpolar domain signals. One approach, known as “hole blowing” involvesidentifying symbols (or samples of symbols) in the baseband signal s(t)during which the magnitude of the signal falls below a predeterminedlow-magnitude threshold, and then raising the magnitude of the basebandsignal s(t) in the temporal vicinity of the identified symbols orsamples so that the AMR of the signal is reduced. The term “holeblowing” is used since the effect of applying the technique is toproduce a “hole” in the signal trajectory diagram of the baseband signals(t). As illustrated in FIG. 7, the “hole” forces the trajectory of themodified baseband signal ŝ(t) to not pass too close to the origin,resulting in a desired reduction in the bandwidth of the signal.

The conventional hole blowing technique is described in U.S. Pat. No.7,054,385 to Booth et al. As explained there, the baseband signal s(t)is modified by adding correction pulses to it, to form a modifiedbaseband signal:

${{\hat{s}(t)} = {{\sum\limits_{n}\;{a_{n}{p\left( {t - {nT}_{s}} \right)}}} + {\sum\limits_{m}\;{b_{m}{r\left( {t - t_{m}} \right)}}}}},$where r(t) is the correction pulse, m is the perturbation index, t_(m)represents the times when the baseband signal s(t) is perturbed (i.e.,the times when the correction pulse r(t) is inserted), and b_(m) is aperturbation sequence representing the amplitude scaling and/or angleshifting applied to the correction pulse r(t).

As shown in FIG. 8, in generating the modified baseband signal ŝ(t) thebaseband signal s(t) from the baseband modulator 102 is fed forward toan analyzer 802. The analyzer 802 then determines the perturbation timest_(m) by detecting low-magnitude events in the baseband signal s(t) thatfall below the fixed low-magnitude threshold. In response to detectedlow-magnitude events, the analyzer 802 generates the perturbationsequence b_(m). A pulse-shaping filter 804 generates the correctionpulses r(t), scales the correction pulses by the perturbation sequenceb_(m), and finally adds the scaled correction pulses to the originalbaseband signal s(t), to produce the desired AMR-reduced modifiedbaseband signal ŝ(t).

The conventional hole blowing technique can be helpful in reducing theAMR of communications signals in polar transmitters configured tooperate in accordance with some types of non-constant envelopemodulation schemes. However, it does not always provide satisfactoryresults. Moreover, for multi-mode polar transmitters that supportmultiple non-constant envelope modulation schemes, which producebaseband signals having different AMRs, the conventional hole blowingtechnique is in most cases unacceptable. State-of-the-art multi-modetransmitters, such as those used in modern cellular telecommunicationssystems, are often designed to operate in both second generation (2G)and third generation (3G) mobile telecommunications systems. 2G mobiletelecommunications systems employ Enhanced Data Rates for GSM (GlobalSystem for Mobile communications) Evolution (EDGE) to achieve higherdata rates. GSM/EDGE uses an 8 phase shift keying (8PSK) non-constantenvelope modulation scheme. 3G Wideband Code Division Multiple Access(W-CDMA) telecommunications systems also use non-constant envelopemodulation schemes—Hybrid Phase Shift Keying (HPSK) and 16 quadratureamplitude modulation (16QAM), if High-Speed Packet Access (HSPA)protocols are used. The baseband signals generated from these differentnon-constant envelope modulation schemes all have different AMRs.Unfortunately, the conventional hole blowing approach described above,with its single, fixed low-magnitude threshold, is inadequate, and inmany cases entirely incapable of, reducing the AMRs of the differentbaseband signals to levels necessary to guarantee compliance with thenoise restriction specifications of all the various standards.

Long Term Evolution (LTE), the next generation mobile telecommunicationstechnology, will also use a non-constant envelope modulation schemeknown as Orthogonal Frequency Division Multiplexing (OFDM). When LTE isdeployed, multi-mode transmitters will be designed to support OFDM aswell as the non-constant envelope modulation schemes used in legacyW-CDMA systems. Problems associated with reducing the AMRs of signals inthose and other next generation multi-mode transmitters will be similarto those encountered in present day GSM/EDGE and W-CDMA multi-modetransmitters.

Considering the drawbacks and limitations of conventional hole blowingapproaches, it would be desirable to have methods and apparatus that areeffective at reducing the AMR of communications signals in current andnext generation communications transmitters.

SUMMARY OF THE INVENTION

Methods and apparatus for reducing the average-to-minimum magnituderatio (AMR) of communications signals in communications transmitters aredisclosed. An exemplary communications transmitter configured to reducethe AMR of a communications signal comprises a symbol mapper, apulse-shaping filter, an AMR reduction circuit, and a modulator (e.g., apolar modulator or quadrature modulator). The symbol mapper operates togenerate a sequence of symbols from a binary-source data streamcontaining a message to be transmitted, and the pulse-shaping filtergenerates a baseband signal based on the sequence of symbols. The AMRreduction circuit is configured to compare magnitudes of samples of thebaseband signal to a time-varying low-magnitude threshold, and modifythe baseband signal to produce a modified baseband signal having areduced AMR if a sample of the baseband signal is determined to have amagnitude less than the time-varying low-magnitude threshold. Finally,the modulator operates to modulate a carrier signal based on themodulation information contained in the modified baseband signal.

In one embodiment of the invention, the AMR reduction circuit includes athreshold setting circuit configured to vary the time-varyinglow-magnitude threshold as a function of one or more of the symbolsgenerated by the symbol mapper (e.g., as a function of the magnitude ormagnitudes of the one or more symbols). In another embodiment, thethreshold setting circuit is configured to set and vary the time-varyinglow-magnitude threshold as a function of one or more samples of thebaseband signal (e.g., as a function of the magnitude or magnitudes ofthe one or more samples). In either embodiment, the function selected todefine the time-varying low-magnitude threshold may comprise a linear ornon-linear function.

The AMR-reducing methods and apparatus of the present invention (alsoreferred to herein as the “adaptive hole blowing” methods and apparatusof the present invention) can be employed in basestation transmitters,cellular handset transmitters, or in any transmitter that can benefitfrom a reduction in the AMR of the transmitter's signals. Further, theAMR-reducing methods and apparatus of the present invention may be usedin transmitters configured to transmit in accordance with a singlenon-constant envelope modulation scheme (i.e., a single-modetransmitter) or in a transmitter that is configurable to transmit inaccordance with multiple non-constant envelope modulation schemes (i.e.,in a multi-mode transmitter).

The AMR-reducing methods and apparatus of the present invention arewell-suited for polar transmitters, to reduce high-frequency events inthe amplitude and angle component signals of the polar transmitter. Thereduction in high-frequency content in the amplitude and angle componentsignals results in lower adjacent channel leakage ratios and lowerreceive band noise. This, together with the ability of the AMR-reducingmethods and apparatus to adapt to the incoming symbols, allows noiserestriction specifications of standards for multiple communicationssystems to be more easily adhered to.

Though well-suited for polar transmitters, the AMR-reducing methods andapparatus of the present invention may also be used in other types oftransmitters, including quadrature-modulator based transmitters.

Further features and advantages of the present invention, includingdescriptions of the structure and operation of the above-summarized andother exemplary embodiments of the invention, will now be described indetail with respect to accompanying drawings, in which like referencenumbers are used to indicate identical or functionally similar elements.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified drawing of a radio frequency (RF) transmitter;

FIG. 2 is a drawing of a conventional quadrature-modulator-basedtransmitter;

FIG. 3 is a drawing illustrating how the modulated RF carrier signalpresented to the RF input port of the power amplifier (PA) of aquadrature-modulator-based transmitter has a non-constant (i.e., timevarying) envelope when the quadrature-modulator-based transmitter isconfigured to operate in accordance with a non-constant envelopemodulation scheme;

FIG. 4 is a drawing of a conventional polar transmitter;

FIG. 5 is a drawing illustrating how the modulated RF carrier signalpresented to the RF input port of the PA of a polar transmitter has aconstant envelope, even when the polar transmitter is configured tooperate in accordance with a non-constant envelope modulation scheme;

FIG. 6 is a signal trajectory diagram of a baseband signal s(t) thatpasses through the origin of the complex signal plane;

FIG. 7 is a signal trajectory diagram of a baseband signal s(t) and amodified baseband signal ŝ(t) that has been subjected to a conventionalhole blowing process;

FIG. 8 is a drawing of an RF transmitter including a conventional holeblowing apparatus;

FIG. 9 is a drawing of a polar transmitter that includes anaverage-to-minimum magnitude ratio (AMR) reduction circuit configured toperform an adaptive hole blowing process, according to an embodiment ofthe present invention;

FIG. 10 is a flowchart of a method performed by the threshold settingcircuit of the AMR reduction circuit of the polar transmitter in FIG. 9to generate a low-magnitude threshold α that varies over time and adaptsto (i.e., is a function of) the symbols generated by the basebandmodulator of the polar transmitter;

FIGS. 11A-F show various types of functions ƒ(S) that may be used todefine the variable/adaptive low-magnitude threshold α=ƒ(S) for the AMRreduction circuit of the polar transmitter in FIG. 9;

FIG. 12 is a graph in the complex signal plane (i.e., the in-phase(I)/quadrature phase (Q) or “I-Q” plane), illustrating how a hole ofradius α=ƒ(S) is formed in the I-Q signal plane and varies continuouslybetween minimum (min) and maximum (max) low-magnitude threshold valuesfor functions ƒ(S) that are continuous;

FIG. 13 is graph of a non-linear step function ƒ(S) having threedifferent threshold levels α1, α2 and α3, which are selected from to setthe low-magnitude threshold α depending on the magnitude (or magnitudes)of one or more symbols generated by the baseband modulator of the polartransmitter in FIG. 9;

FIG. 14 is a graph in I-Q signal plane, illustrating how three differentholes of radiuses α1, α2 and α3 are formed in the I-Q signal plane whenthe low-magnitude threshold α=ƒ(S) is a three-step step function, likethat in FIG. 13;

FIG. 15 is a drawing of a threshold setting circuit that may be used toset the low-magnitude threshold α=ƒ(S) in the AMR reduction circuit ofthe polar transmitter in FIG. 9;

FIG. 16 is a flowchart of an adaptive hole blowing method performed bythe AMR reduction circuit of the polar transmitter in FIG. 9, accordingto an embodiment of the present invention;

FIG. 17 is a vector diagram in the I-Q signal plane illustrating how theadaptive hole blowing method in FIG. 16 determines which sample among athree-sample set (I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1), Q_(i−1))has the lowest magnitude;

FIG. 18 is a vector diagram in the I-Q signal plane illustrating how theadaptive hole blowing method in FIG. 16 determines which sample amongthe three-sample set (I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1),Q_(i−1)) has the next-lowest magnitude;

FIG. 19A is a schematic of a local minimum event detection circuit thatmay be used to detect local minimum events during performance of theadaptive hole blowing method in FIG. 16;

FIG. 19B is a schematic of an alternative local minimum event detectioncircuit that may be used to detect local minimum events duringperformance of the adaptive hole blowing method in FIG. 16;

FIG. 20 is a vector diagram in the I-Q signal plane of a trajectoryvector (Δx, Δy), which is formed during performance of the adaptive holeblowing method in FIG. 16;

FIG. 21A is a schematic of a trajectory vector calculation circuit thatmay be used to generate the trajectory vector (Δx, Δy) shown in FIG. 20during performance of the adaptive hole blowing method in FIG. 16;

FIG. 21B is a schematic of an alternative trajectory vector calculationcircuit that may be used to generate the trajectory vector (Δx, Δy)shown in FIG. 20 during performance of the adaptive hole blowing methodin FIG. 16;

FIG. 22 is a vector diagram in the I-Q signal plane of a firstorthogonal vector (Δy, −Δx) and an opposing second orthogonal vector(−Δy, Δx), which are formed during performance of the adaptive holeblowing method in FIG. 16;

FIG. 23 is a vector diagram in the I-Q signal plane illustrating how theappropriate one of the first and second orthogonal vectors (Δy, −Δx) and(−Δy, Δx) in FIG. 22 is selected during performance of the adaptive holeblowing method in FIG. 16;

FIG. 24 is a schematic of a circuit that may be used to select theappropriate one of the first and second orthogonal vectors (Δy, −Δx) and(−Δy, Δx) shown in FIG. 22 during performance of the adaptive holeblowing method in FIG. 16;

FIG. 25 is a vector diagram in the I-Q signal plane illustrating thelocation of a threshold sample (I_(c), Q_(c)) on the low-magnitudethreshold circle α=ƒ(S) during performance of the adaptive hole blowingmethod in FIG. 16;

FIG. 26 is a schematic of a pulse insertion vector generator circuitthat may be used to form a pulse insertion vector (Ins_I, Ins_Q) duringperformance of the adaptive hole blowing method in FIG. 16;

FIG. 27 is a vector diagram in the I-Q signal plane of the pulseinsertion vector (Ins_I, Ins_Q) generated by the pulse insertion vectorgenerator circuit in FIG. 26, and the relationship of the pulseinsertion vector (Ins_I, Ins_Q) to the trajectory vector (Δx, Δy) andthe sample vector (I_(i), Q_(i));

FIG. 28 is a schematic of an alternative pulse insertion vectorgenerator circuit that may be used to form the pulse insertion vector(Ins_I, Ins_Q) during performance of the adaptive hole blowing method inFIG. 16;

FIG. 29A is a signal trajectory diagram obtained from simulationsperformed on a polar transmitter, similar to the polar transmitter inFIG. 9, in which the polar transmitter was configured to process andtransmit a representative High-Speed Uplink Packet Access (HSUPA) signaland the AMR reduction circuit was configured to perform adaptive holeblowing using a tri-level (α2, α2, α3) low-magnitude threshold α definedby a three-step step function ƒ(S), similar to that shown in FIG. 13;

FIG. 29B is a signal trajectory diagram obtained from simulationsperformed on a polar transmitter configured to process and transmit thesame representative HSUPA signal used in the simulations performed inconnection with FIG. 29A, but with the polar transmitter configured toperform hole blowing using a single, fixed low-magnitude threshold;

FIG. 30 is a diagram of a polar modulator that includes a phasemodulation (PM)-path non-linear phase filter, an amplitude modulation(AM)-path linear finite impulse response (FIR) filter, and/or a PM-pathlinear FIR filter, and which may be substituted for the polar modulatorof the polar transmitter in FIG. 9, in accordance with variousembodiments of the present invention;

FIG. 31 is a table comparing the adjacent channel leakage ratio (ACLR),receive band noise (R×N), and error vector magnitude (EVM) performanceof various embodiments of the invention using the adaptive hole blowingaspect of the present invention to the ACLR, R×N and EVM performance ofa polar transmitter configured to perform hole blowing using a single,fixed low-magnitude threshold; and

FIG. 32 is a drawing of a quadrature-modulator-based transmitter that isadapted to include an AMR reduction circuit configured to perform anadaptive hole blowing method, similar to the adaptive hole blowingmethod shown in FIG. 16, according to an embodiment of the presentinvention.

DETAILED DESCRIPTION

Referring to FIG. 9, there is shown a polar transmitter 900, accordingto an embodiment of the present invention. The polar transmitter 900 maybe used in a variety of different applications, including in an accesspoint or basestation, or in a mobile handset, such as is used in amobile telecommunications network, for example. The polar transmitter900 comprises a baseband modulator 902, an average-to-minimum magnituderatio (AMR) reduction circuit 904, and a polar modulator 906. Thebaseband modulator 902 and AMR reduction circuit 904 are implemented asa digital signal processor (DSP) comprised of hardware or a combinationof hardware and software, such as a microprocessor, microcontroller,field-programmable gate array, or other programmable or nonprogrammableintegrated circuit, and are formed in one or more digital integratedcircuits, as will be appreciated by those of ordinary skill in the art.

The baseband modulator 902 includes a symbol mapper 908 and apulse-shaping filter 910. The symbol mapper 908 is configured togenerate a sequence of symbols comprised of in-phase andquadrature-phase components I′(t) and Q′(t) from a binary-source datastream containing a digital message to be transmitted. The polartransmitter 900 comprises either a single-mode polar transmitter or amulti-mode transmitter. Accordingly, the symbol mapper 908 is configuredto operate according to either a single modulation scheme or inaccordance with multiple modulation schemes. According to oneembodiment, the symbol mapper 902 is configured to format the sequenceof symbols according to the Hybrid Phase Shift Keying (HPSK) modulationscheme specified by the Third Generation Partnership Project (3GPP) foruse in the Wideband Code Division Multiple Access (W-CDMA) air interfaceof Universal Mobile Telecommunications System (UMTS) networks andaccording to the Quadrature Phase Shift Keying (QPSK) and 16 QuadratureAmplitude Modulation (16QAM) modulation schemes specified by the 3GPPfor use in UMTS networks supporting the High-Speed Packet Access (HSPA)communication protocols.

The pulse-shaping filter 910, which in one embodiment comprises aroot-raised-cosine filter, is configured to pulse shape the I′(t) andQ′(t) components of the sequence of symbols from the symbol mapper 908and then sample the result, to provide digital samples of a complexbaseband signal s(t) having in-phase and quadrature-phase componentsQ(t) and I(t).

The AMR reduction circuit 904 comprises a threshold setting circuit 912,a main signal path that includes a delay element 914, I and Q summers916 and 918, and Coordinate Rotation Digital Computer (CORDIC) converter920; and a feed-forward path that includes a local minimum detector 922,pulse insertion vector generator 924, pulse generator 926, and I and Qmultipliers 928 and 930.

The threshold setting circuit 912 is configured to generate alow-magnitude threshold α that adapts to (or is a function of) thesymbols generated by the symbol mapper 908. The local minimum detector922 is configured to receive the low-magnitude threshold α and comparesamples of the digital baseband signal components I(t) and Q(t) to thelow-magnitude threshold α. If the local minimum detector 922 detects alow-magnitude event, i.e., detects that the magnitude (I²+Q²)^(1/2) of agiven sample of the baseband signal s(t) is less than the low-magnitudethreshold α, it generates a local minimum event detected (LMED) signal,and the pulse insertion vector generator 924 generates a pulse insertionvector (Ins_I, Ins_Q).

The pulse insertion vector (Ins_I, Ins_Q) is, in general, complex,having a magnitude and phase that affect how the amplitude and/or angleof the baseband signal s(t) is/are to be modified to remove thelow-magnitude event. The baseband signal s(t) is modified by a scaledinsertion pulse (p_(I)t), p_(Q)(t)), which is generated by scaling apulse (using the I and Q multipliers 928 and 930) provided by the pulsegenerator 926 by the pulse insertion vector (Ins_I, Ins_Q). The scaledinsertion pulse (p_(I)t), p_(Q)(t)) is combined with the I(t) and Q(t)components of the baseband signal in the main signal path of the AMRreduction circuit 904 via the I and Q summers 916 and 918. The delayelement 914 is set so that the scaled insertion pulse (p_(I)(t),p_(Q)(t)) is properly combined with the baseband signal s(t) in thetemporal vicinity of the detected low-magnitude event.

According to one embodiment of the invention, the threshold settingcircuit 912 is configured to generate a low-magnitude threshold a thatvaries with time and is a function of the magnitudes of the symbolsgenerated by the symbol mapper 908. In other words, α=ƒ(S).

An exemplary method 1000 of setting the low-magnitude threshold αaccording to this embodiment of the invention is shown in FIG. 10. In afirst step 1002 of the method 1000, a first set of symbols S(n), . . . ,S(n+m), where m≧0, is loaded into the threshold setting circuit 912.Next, at step 1004 the low-magnitude threshold α is computed accordingto a predefined function ƒ(S) and as a function of the magnitude (ormagnitudes) of one or more of the symbols in the first set of symbolsS(n), . . . , S(n+m). The magnitude (or magnitudes) of the one or moresymbols can be based on the minimum, maximum or an average of themagnitudes of one or more symbols in the first set of symbols S(n), . .. , S(n+m).

At step 1006 the threshold is limited to minimum (min) and maximum (max)values. Then, at step 1008 the value of the min/max limitedlow-magnitude threshold α is sent to the local minimum detector 922,which, as described above, operates to compare the magnitudes of samplesof the digital baseband signal s(t) to the low-magnitude threshold α.

The local minimum detector 922 continues comparing the magnitudes of thesamples of the digital baseband signal s(t) to the low-magnitudethreshold α until at decision 1010 it is determined that a newlow-magnitude threshold α is to be calculated and used. When thiscondition occurs, the next set of symbols S(n+1), . . . , S(n+m+1) isloaded into the threshold setting circuit 912 and steps 1002-1010 arerepeated.

It should be mentioned here that, while the threshold setting circuit912 in this exemplary embodiment is configured to generate thelow-magnitude threshold α as a function of the magnitudes of theincoming symbols (i.e., prior to the pulse-shaping filter 910, as inFIG. 9), it can be alternatively configured to generate thelow-magnitude threshold α as a function of the magnitudes of samples (oroversamples, if oversampling is used) of the digital baseband signals(t) produced at the output of the pulse shaping filter 910.

The function ƒ(S) defining the low-magnitude threshold α can be acontinuous function (linear or polynomial) or a non-continuous (i.e.,nonlinear) function. FIGS. 11A-F show various functions that may beused. When the polar transmitter 900 is configured for multi-modeoperation, the same or different functions may be used for the variousmodulation schemes that the multi-mode polar transmitter is configurableto operate.

When plotted in the complex (I-Q) signal plane, the hole that isproduced in the signal trajectory has a radius equal to α, and the size(i.e., radius) of the hole varies between the minimum (min) and maximum(max) low-magnitude threshold values. When the selected function ƒ(S) isa linear or polynomial function, the hole size varies continuouslybetween the min and max values, as illustrated in the I-Q signal planediagram in FIG. 12. When the selected function ƒ(S) is non-linear, likethe step function in FIG. 13, a plurality of holes of different sizesare generated, as illustrated in FIG. 14.

The rate at which the hole size varies depends on the rate at which theincoming symbols are examined by the threshold setting circuit 912. Inone embodiment of the invention, the rate at which the incoming symbolsare examined corresponds to the 3.84 Mchips/sec chip rate used in anUMTS system. However, the rate can be set to other values, includingmultiples or fractions of the chip rate, or multiples or fractions ofthe sampling or oversampling rate used in generating the I and Qcomponent samples of the complex baseband signal s(t).

In one embodiment of the invention, the three-step step function ƒ(S) inFIG. 13 is used, m, which establishes the size of the sets of symbolsS(n), . . . , S(n+m), is set equal to one (i.e., m=1) to form sets ofsymbols with two symbols each, and the threshold setting circuit 912 isconfigured to set the low-magnitude threshold a to α1, α2 or α3,depending on the magnitude of the lowest-magnitude symbol in each twosymbol set.

For each two symbol set that is examined, the threshold setting circuit912 sets the low-magnitude threshold α to α1, α2 or α3. For a two symbolset in which the magnitude of the lowest-magnitude symbol is less thanL1 (refer to FIG. 13), the threshold setting circuit 912 sets α equal toα1. For a two symbol set in which the magnitude of the lowest-magnitudesymbol is greater than or equal to L1 but less than L2, the thresholdsetting circuit 912 sets a equal to α2. And for a two symbol set inwhich the magnitude of the lowest-magnitude symbol is greater than orequal to L2, the threshold setting circuit 912 sets α equal to α3.

Once the low-magnitude threshold a has been set to α1, α2 or α3, thelocal minimum detector 922 operates to detect local minimum events inthe baseband signal s(t) using that value of α. The same value of α isused until the threshold setting circuit 912 examines symbols in asubsequent two symbol set requiring a change of the low-magnitudethreshold a from its present level to a different one of the thresholdlevels α1, α2 or α3.

The actual values of the lower and upper levels L1 and L2 and thethreshold levels α1, α2 and α3 are determined beforehand by test orsimulation. In one embodiment, where the symbol mapper 902 is configuredto format the sequences of symbols I′(t) and Q′(t) according to bothHPSK and High-Speed Uplink Packet Access (HSUPA), the min, mid and maxthreshold levels a2, a2 and a3 and lower and upper levels L1 and L2 areset so that a desired combination of in-band and out-of-band noiseperformance characteristics are satisfied for a plurality of differentHPSK and HSUPA signals the polar transmitter 900 is configurable totransmit.

FIG. 15 is a drawing of a threshold setting circuit 1500 that may beused to implement the threshold setting circuit 912 in FIG. 9, when thefunction ƒ(S) defining the low-magnitude threshold α is a three-stepstep function ƒ(S), as in FIG. 13. The threshold setting circuit 1500operates to determine which symbol of two consecutive symbols S₁=(I′₁,Q′₁) and S₂=(I′₂, Q′₂) has the lowest magnitude, and based on theresult, sets the low-magnitude threshold a to one of 2-bit thresholdlevels α1, α2 or α3. More specifically, first and second summers 1502and 1504 first determine which of the I and Q components in each of thesymbols S₁ and S₂ has the lowest magnitude. The I and Q components arerepresented in two's complement form with the most significant bit (MSB)operating as a sign bit. Accordingly, the first and second summers 1502and 1504 operate as comparators, the sign bit at the output of the firstsummer 1502 indicating which of the I and Q components of the firstsymbol S₁ is of the lowest magnitude and the sign bit at the output ofthe second summer 1502 indicating which of the I and Q components of thesecond symbol S₂ has the lowest magnitude.

The sign bits at the outputs of the first and second summers 1502 and1504 are coupled to the select inputs of first and second multiplexers1506 and 1508, so that the I or Q component of the first symbol S₁having the lowest magnitude is produced at the output of the firstmultiplexer 1506 and the I or Q component of the second symbol S₂ havingthe lowest magnitude is produced at the output of the second multiplexer1508. A third summer 1510 and third multiplexer 1512 operate similarly,selecting the lowest-magnitude output from among the two outputs of thefirst and second multiplexers 1506 and 1508. Finally, depending onmagnitude of the output of the third multiplexer 1512, decoding logic1514 sets the low-magnitude threshold a to the appropriate 2-bitthreshold level α1, α2 or α3.

The threshold setting circuit 1500 operates on the basis that themagnitudes of the symbols S₁ and S₂ are constrained to a limited numberof known values (defined by the particular modulation scheme beingemployed), and the ability to determine the magnitude of thelowest-magnitude symbol by only having to determine the magnitude of thelowest-magnitude I or Q component in the two symbols S₁ and S₂. Themagnitudes of the symbols themselves need not be directly computed,thereby avoiding the need for computationally intensive square rootcalculations. Nevertheless, other threshold setting circuits thatcompute the magnitudes of the symbols S₁ and S₂ (either directly orindirectly) and which set the low-magnitude threshold a based on acomparison of the computed results may be alternatively used.

FIG. 16 is a flowchart of an adaptive hole blowing method 1600 thatincorporates the variable/adaptive low-magnitude threshold aspect of thepresent invention described above. In a first step 1602 of the method1600 a first three-sample set of samples (I_(i+1), Q_(i+1)), (I_(i),Q_(i)), (I_(i−1), Q_(i−1)) is loaded into the local minimum event 922,where i is a sample index and (I_(i+1), Q_(i+1)), (I_(i), Q_(i)) and(I_(i−1), Q_(i−1)) are referred to in the description that follows asthe next, middle and prior samples, respectively. In one embodiment, thenext, middle and prior samples (I_(i+1), Q_(i+1)), (I_(i), Q_(i)),(I_(i−1), Q_(i−1)), and samples in subsequent sample sets, aretemporally adjacent (i.e., are consecutive). However, that not need bethe case, and in an alternative embodiment are temporally nonadjacent(i.e., are nonconsecutive).

In steps 1604 and 1606 (see FIGS. 17 and 18) the local minimum detector922 determines which sample among the samples in the first three-sampleset (I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1), Q_(i−1)) has thelowest magnitude (step 1604), and which of the samples has thenext-lowest magnitude (step 1606).

Next, at decision 1608, the local minimum detector 922 determineswhether a local minimum event is present in the first three-sample set(I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1), Q_(i−1)). The process ofdetecting a local minimum event can be performed in various way. In theexemplary embodiment described here, a local minimum event is deemedpresent if the middle sample (I_(i), Q_(i)) in the three-sample set(I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1), Q_(i−1)) has a magnitudethat is less then the current value of the low-magnitude threshold α andis the sample among the three samples with the lowest magnitude.

If a local minimum event is not detected (“no” at decision 1608), atstep 1624 the next three-sample set (I_(i+2), Q_(i+1)), (I_(i+1),Q_(i+1)), (I_(i), Q_(i)) is loaded into the local minimum detector 922and steps 1604-1608 are repeated. On the other hand, if a local minimumevent is detected (“yes” at decision 1608), the local minimum detector922 generates an LMED (local minimum event detected) output signal,which is used to signal the pulse insertion vector generator 924 togenerate a non-zero pulse insertion vector (Ins_I, Ins_Q), as explainedin more detail below.

FIG. 19A is a drawing of a local minimum event detection circuit 1900that may be used to implement the local minimum event detection portionof the local minimum detector 922. The local minimum event detectioncircuit 1900 comprises a group of multipliers 1902, a first group ofadders 1904, a second group of adders 1906, and a NOR logic gate 1908.The multipliers of the group of multipliers 1902 and the adders of thefirst and second groups of adders 1904 and 1906 may be formed in avariety of different ways. For example, the multipliers may be formedfrom logic gates using Wallace trees or as Dadda multipliers and theadders may be formed from logic gates using ripple-carry orcarry-lookahead adders, as will be appreciated and understood by thoseof ordinary skill in the art.

The group of multipliers 1902 and the first group of adders 1904 operateto form the sums of the squares of the I and Q components of each sampleof the three-sample set (I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1),Q_(i−1)), i.e., M_(i+1)=[(I_(i+1))²+(Q_(i+1))²].M_(i)=[(I_(i))²+(Q_(i))²] and M_(i−1)=[(I_(i−1))²+(Q_(i−1))²].Collectively, the sums of the squares M_(i+1), M_(i) and M_(i−1) providean accurate indication of the relative magnitudes of the three samples(I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1), Q_(i−1)).

The second group of adders 1906 operates to subtract the square of themagnitude of the middle sample (I_(i), Q_(i)) from the square of themagnitude of the next sample (I_(i+1), Q_(i+1)), and also subtract thesquare of the magnitude of the middle sample (I_(i), Q_(i)) from thesquare of the magnitude of the prior sample (I_(i−1), Q_(i−1)). The MSBsign bits at the outputs of the second group of adders 1906 determinewhether the magnitude of the middle sample (I_(i), Q_(i)) is the lowestmagnitude among the prior, middle and next samples. If it is, the MSBsign bits at the outputs of both adders of the second group of adders1906 are both at a logic “0” and the LMED output of the NOR logic gate1908 is a logic “1,” indicating the detection of a local minimum event.Otherwise, the LMED output remains at a logic “0.”

FIG. 19B is a drawing of an alternative local minimum event detectioncircuit 1910 that may be used to implement the local minimum eventdetection portion of the local minimum detector 922. The alternativelocal minimum event detection circuit 1910 is similar to the localminimum event detection circuit 1900 in FIG. 19A, except that itincorporates delay (“D”) flip-flops 1912 and 1914. The D flip-flops 1912and 1914 provide a pipelining function for the samples, thereby reducingthe number of multipliers and adders that are needed to perform thelocal minimum event detection.

If the local minimum detector 922 has detected the occurrence of a localminimum event at decision 1608 of the method 1600, at step 1610 atrajectory vector (Ax, Δy) approximating the trajectory of the basebandsignal s(t) through the three samples (I_(i+1), Q_(i+1)), (I_(i) Q_(i)),(I_(i−1), Q_(i−1)) is calculated. According to one embodiment,illustrated in FIG. 20, the trajectory vector (Δx, Δy) is defined as thevector (Min1−Min 2)=[(I_(i)−I_(i−1)), (Q_(i)−Q_(i−1))]=(Δx, Δy) betweenthe lowest-magnitude sample (I_(i−1), Q_(i−1)) and thenext-lowest-magnitude sample (in this example, the prior sample(I_(i−1), Q_(i−1)) is the next-lowest-magnitude sample). In analternative embodiment, the trajectory vector (Δx, Δy) is defined as thevector between the prior and next samples, i.e., (Δx,Δy)=[(I_(i+1)−I_(i−1)), (Q_(i+1)−Q_(i−1))].

FIG. 21A is a drawing of a trajectory vector calculation circuit 2100that may be used to compute the trajectory vector (Δx, Δy) in step 1610of the adaptive hole blowing method 1600. The trajectory vectorcalculation circuit 2100 comprises a group of multipliers 2102, a groupof adders 2104, an MSB sign bit adder 2106, first and secondmultiplexers 2108 and 2110, and first and second output adders 2112 and2114. The group of multipliers 2102 and group of adders 2104 operate todetermine the squares of the magnitudes of the prior and next samples(I_(i−1), Q_(i−1)) and (I_(i+1), Q_(i+1)). The MSB sign bit adder 2106subtracts the square of the magnitude of the next sample (I_(i+1),Q_(i+1)) from the square of the magnitude of the prior sample (I_(i−1),Q_(i−1)). The MSB sign bit at the output of the MSB sign bit adder 2106provides an indication of which of the prior and next samples (I_(i−1),Q_(i−1)) and (I_(i+1), Q_(i+1)) has the lowest magnitude. The one thathas the lowest magnitude is the sample that has the next-lowestmagnitude among the samples of the three-sample set (I_(i+1), Q_(i+1)),(I_(i), Q_(i)), (I_(i−1), Q_(i−1)).

The MSB sign bit formed at the output of MSB sign bit adder 2106 isdirected to the select inputs of the first and second multiplexers 2108and 2110. Accordingly, if the MSB sign bit has a value indicating thatthe next sample (I_(i+1), Q_(i+1)) is the next-lowest-magnitude sampleamong the three-sample set (I_(i+1), Q_(i+1)), (I_(i), Q_(i)), (I_(i−1),Q_(i−1)), the I and Q components of the next sample (I_(i+1), Q_(i+1))are passed to the outputs of the first and second multiplexers 2108 and2110. Otherwise, the I and Q components of the prior sample (I_(i−1),Q_(i−1)) are passed to the multiplexer outputs.

Finally, the first and second output adders 2112 and 2114 subtract the Iand Q components of the middle sample (I_(i), Q_(i)) from the outputs ofthe first and second multiplexer 2108 and 2110 to produce the trajectoryvector (Δx, Δy).

As explained above, in an alternative embodiment, the trajectory vector(Δx, Δy) is defined by the vector difference between the next and priorsamples, i.e., (Δx, Δy)=[(I_(i+1)−I_(i−1)), (Q_(i+1)−Q_(i−1))]. FIG. 21Bis a trajectory vector calculation circuit 2120 that may be used togenerate the trajectory vector (Δx, Δy) according to that alternativeembodiment.

At step 1612 the orthogonal vector (i.e., the vector that is orthogonalto the trajectory vector) is determined. There are two possibleorthogonal vectors, a first orthogonal vector (Δy, −Δx) and an opposingsecond orthogonal vector (−Δy, Δx), as illustrated in FIG. 22. To ensureproper AMR reduction, it is necessary to select the orthogonal vectorhaving the appropriate direction. The orthogonal vector that has theappropriate direction is the one that facilitates pushing the signaltrajectory of the baseband signal s(t) away from the origin, rather thantoward it. This selection process, which is performed in step 1614, canbe performed in a variety of different ways. An exemplary orthogonaldirection selection algorithm is described below.

(1) Solve for (x′, y′) orthogonal direction from intersection oftrajectory and orthogonal vectors (see FIGS. 22 and 23): i. by − ax = cbasic formula ii. Q_(i)Δx − I_(i)Δy = c equation for trajectory vectorthrough (I_(i), Q_(i)) iii. y′Δx − x′Δy = c equation for trajectoryvector through (x′, y′) iv. Q_(i)Δx − I_(i)Δy = y′Δx − x′Δy substitutefor constant c v. y′ = x′Δx/Δy equation for orthogonal vector through(x′, y′) vi. Q_(i)Δx − I_(i)Δy = (x′Δx/Δy)Δx − x′Δy substitute for y′ inequation vi and solve for x′ vii. −ΔyQ_(i)Δx − I_(i)Δy) = (Δx² + Δy²)x′(Δx² + Δy²) is positive and can be dropped viii. −ΔyQ_(i)Δx − I_(i)Δy) =x′ ix. Q_(i)Δx − I_(i)Δy = y′Δx − (y′Δy/Δx)Δy substitute for x′ inequation vi and solve for y′ x. Δx(Q_(i)Δx − I_(i)Δy) = (Δx² + Δy²)y′(Δx² + Δy²) is positive and can be dropped xi. Δx(Q_(i)Δx − I_(i)Δy) =y′ (2) Sign of (x′, y′) used to find appropriate direction of orthogonalvector.

In the example shown and described here, the first orthogonal vector(Δy, −Δx) is determined and selected to be the appropriate orthogonalvector. The selected orthogonal vector (Δy, −Δx) and its direction areshown in FIG. 23. After the appropriate orthogonal vector direction isdetermined at step 1614, at step 1616 the coordinates of the selectedorthogonal vector are used as a reference into a lookup table (LUT) toretrieve a threshold sample (I_(c), Q_(c)) that intersects with thelow-magnitude threshold circle α and the selected orthogonal vector (Δy,−Δx), as illustrated in FIG. 24. Because the low-magnitude threshold αis a function of the incoming symbols, i.e., α=ƒ(S), the thresholdsample (I_(c), Q_(c)) varies depending on the current value of thelow-magnitude threshold a provided by the threshold setting circuit 912.In the exemplary described above, where ƒ(S) is a three-step stepfunction (see FIGS. 13 and 14 and accompanying description) threeseparate pairs of sine and cosine LUTs are used, as illustrated in FIG.26, each LUT corresponding to one of the three different possiblethreshold levels α1, a2 and a3. Which pair of LUTs is used depends onwhich of the three threshold levels α1, α2 and α3 the low-magnitudethreshold α is set to (as provided by the threshold setting circuit1500).

At step 1618 the I and Q components of the lowest-magnitude middlesample (I_(i), Q_(i)) are subtracted from the I and Q components of thethreshold sample (I_(c), Q_(c)) to determine three different pulseinsertion vectors (Ins_I, Ins_Q), each corresponding to one of the threedifferent threshold levels α1, α2 and α3. As shown in FIG. 26, theappropriate threshold-dependent pulse insertion vector (Ins_I, Ins_Q) isselected at the outputs of first and second pulse insertion vectormultiplexers 2602 and 2604 using the current value of the low-magnitudethreshold α as the select input for each of the multiplexers 2602 and2604. Note that if a low-magnitude event was not previously detected bythe local minimum detector 922, the LMED signal has αvalue that switchesthe inputs to the first and second pulse insertion vector multiplexers2602 and 2604 to zero, effectively providing a pulse insertion vector(Ins_I, Ins_Q) of zero value to the multiplexer outputs. The pulseinsertion vector (Ins_I, Ins_Q) and its relationship to the trajectoryvector (Δx, Δy) and the sample vector (I_(i), Q_(i)) are shown in FIG.27.

Once the pulse insertion vector (Ins_I, Ins_Q) has been determined, atstep 1620 the I and Q multipliers 928 and 930 scale a complex insertionpulse provided by the pulse generator 926 by the pulse insertion vectors(Ins_I, Ins_Q), to provide the desired, scaled complex insertion pulse(p_(I)t), p_(Q)(t)). The impulse response of the pulse generator 926 canbe the same as the pulse shaping filter 910 or may be different. In oneembodiment, the impulse response of the pulse generator 926 issubstantially similar to the impulse response of a root-raised-cosinefilter. However, it may have different shapes, e.g., Gaussian,triangular, etc. After the scaled complex insertion pulse (p_(I)t),p_(Q)(t)) is generated, it is added to the baseband signal s(t) in themain signal path of the AMR reduction circuit 904 (via the I and Qsummers 916 and 918 in FIG. 9) in the temporal vicinity of thelowest-magnitude middle sample (I_(i), Q_(i)), thereby reducing the AMRof the baseband signal s(t).

Finally, at decision 1622 it is determined whether the threshold settingcircuit 912 is ready to set the low-magnitude threshold a to a newvalue. If it is (“yes”), the threshold setting circuit 912 performsthreshold setting method 1000 (see FIG. 10) on a next set of symbolsbefore continuing with the adaptive hole blowing method 1600. Once themethod 1000 is completed, the adaptive hole blowing method 1600 resumesat step 1624, where the next set of samples (I_(i+2), Q_(i+2)),(I_(i+1), Q_(i+1)), (I_(i), Q_(i)) is loaded for examination by thelocal minimum detector 922 using the new value of the low-magnitudethreshold α. If, on the other hand at decision 1622 it is determinedthat a new low-magnitude threshold α does not need to be adjusted to anew level, the adaptive hole blowing method 1600 proceeds directly tostep 1624 (i.e., without first performing the threshold setting method1000 in FIG. 10), and the next set of samples (I_(i+2), Q_(i+2)),(I_(i+1), Q_(i+1)), (I_(i), Q_(i)) is loaded for examination by thelocal minimum detector 922.

In the circuit in FIG. 26, a plurality LUTs are used to store the sineand cosine information defining the threshold sample (I_(c), Q_(c)). Thethreshold sample from the appropriate pair of sine/cosine LUTs is thenused to generate the desired pulse insertion vector (Ins_I, Ins_Q). Inan alternative embodiment shown in FIG. 28, which does not require theuse of sine/cosine LUTs, the pulse insertion vector (Ins_I, Ins_Q) isbased directly on the selected orthogonal vector (Δy, −Δx).Specifically, the pulse insertion vector (Ins_I, Ins_Q) is assigned thesame direction as the selected orthogonal vector (Δy, −Δx) and amagnitude that is one of three predefined fractions (in this example, ¼,½ and 1) of the selected orthogonal vector (Δy, −Δx) magnitude. Which ofthe three different magnitudes is selected is determined by whichthreshold level, α1, α2 or α3 the low-magnitude threshold α is set to.This approach obviates the need for the sine/cosine LUTs. Not requiringthe sine/cosine LUTs can be beneficial if, for example, the read onlymemory needed to implement the LUTs is not readily available or ifintegrated circuit size constraints prevent or hinder integration orplacement of the LUTs.

It should be emphasized that the methods and apparatus for generatingthe pulse insertion vector (Ins_I, Ins_Q) and forming the insertionpulse (p_(I)t), p_(Q)(t)), once the low-magnitude threshold a has beenset for a particular set of symbols, are merely exemplary. Other methodsand apparatus for generating the pulse insertion vector (Ins_I, Ins_Q)and forming the insertion pulse (p_(I)t), p_(Q)(t)), which with thebenefit of this disclosure may be modified for use with theadaptive/variable low-magnitude threshold aspect of the presentinvention, are described in commonly assigned U.S. patent applicationSer. Nos. 12/414,016, 12/482,913 and 12/508,477, all of which are herebyincorporated by reference.

Modifying the baseband signal s(t)=I(t)+jQ(t) by the scaled insertionpulses (p_(I)(t), p_(Q)(t)) results in a modified rectangular-coordinatebaseband signal ŝ(t)=Î (t)+j{circumflex over (Q)}(t) having a reducedAMR. The CORDIC converter 920 then converts this modifiedrectangular-coordinate baseband signal ŝ(t)=Î(t)+j{circumflex over(Q)}(t) to polar-coordinate amplitude and angle component signals{circumflex over (ρ)}(t) and {circumflex over (θ)}(t), as shown in FIG.9. Due to the prior AMR-reducing operation performed by the AMRreduction circuit 904, the polar-coordinate amplitude and anglecomponent signals {circumflex over (ρ)}(t) and {circumflex over (θ)}(t)have reduced high-frequency content.

To complete the polar modulation and upconversion process and render amodulated RF carrier signal suitable for transmission over the air to aremote receiver, the polar-coordinate amplitude and angle componentsignals {circumflex over (ρ)}(t) and {circumflex over (θ)}(t) from theAMR reduction circuit 904 are coupled to the amplitude modulation (AM)and phase modulation (PM) paths of the polar modulator 906. The AM pathof the polar modulator 906 includes an AM path digital-to-analogconverter (DAC) 932 an AM path analog low-pass filter (LPF) 934 and anamplitude modulator 936. The PM path includes a PM path DAC 938, a PMpath analog filter 940, and an angle modulator 942. The AM and PM pathDACs 932 and 938 convert the amplitude and angle component signals{circumflex over (ρ)}(t) and {circumflex over (θ)}(t) to analogamplitude and phase modulation signals, respectively. The AM and PM pathanalog LPFs 934 and 940 then filter out undesirable DAC images createdduring the digital-to-analog conversion process. The resulting filteredamplitude modulation signal is fed to the amplitude modulator 936, whilethe resulting filtered angle modulation signal is fed to the anglemodulator 942.

The amplitude modulator 936 modulates a direct current power supplyvoltage Vsupply according to the amplitude information in the filteredamplitude modulation signal. The resulting amplitude-modulated powersupply signal Vs(t) is coupled to the power supply port of the polarmodulator's power amplifier (PA) 944. Meanwhile, in the PM path theangle modulator 942 operates to modulate an RF carrier signal accordingto the angle information in the filtered angle modulation signal. Theresulting angle-modulated RF carrier signal is applied to the RF inputport RFin of the PA 944.

The PA 944 comprises an amplifier having a final-stage switch-mode typeof PA (e.g., a Class D, E or F switch-mode PA) operating betweencompressed and cut-off states. As the PA 944 amplifies theangle-modulated RF carrier signal produced at the output of the anglemodulator 942, the envelope information in the amplitude-modulated powersupply signal Vs(t) from the amplitude modulator 936 is restored at theRF output RFout of the PA 944. Finally, an antenna 946 radiates theresulting amplified amplitude- and angle-modulated RF carrier signalover the air to a remote receiver.

FIG. 29A is a signal trajectory diagram obtained from simulationsperformed on a polar transmitter, similar to the polar transmitter 900in FIG. 9, in which the polar transmitter 900 was configured to processand transmit a representative HSUPA signal and the AMR reduction circuit904 was configured to perform adaptive hole blowing using a tri-level(α2, α2, α3) low-magnitude threshold a defined by a three-step stepfunction ƒ(S), similar to that shown in FIG. 13. FIG. 29B is a signaltrajectory diagram obtained from simulations performed on a polartransmitter configured to process and transmit the same representativeHSUPA signal, but with the polar transmitter using a single, fixedlow-magnitude threshold. Although faint, the three different hole sizesof radiuses of α1, α2 and α3 can be seen in the signal trajectorydiagram in FIG. 29A. Simulation data on the two different configurationshas also shown that the adaptive hole blowing approach of the presentinvention yields similar or lower upper and lower first adjacent (+/−5MHz offset from uplink channel) channel leakage ratios (ACLR5s) comparedto the fixed threshold approach, significantly lower upper and lowersecond ACLR10s (+/−10 MHz offsets from uplink channel), significantlylower receive band noise (R×N) at a 45 MHz frequency offset and higher,and similar or better in-band error vector magnitude (EVM) performance.

In accordance with other embodiments of the invention, further ACLRreduction (both first and second adjacent channels) and further R×Nreduction are achieved by including one or more of the following in theAM and PM paths of the polar modulator 906: a PM-path non-linear phasefilter 3006, an AM-path linear finite impulse response (FIR) filter3008, and a PM-path linear FIR filter 3010. The inclusion of thesevarious filters 3006, 3008 and 3010 is illustrated in the modified polarmodulator 3004 in FIG. 30, where the dashed boxes are used to indicatethat the filters are optional.

FIG. 31 is a table comparing the ACLR5s, ACLR10s, R×N45 and EVM of apolar transmitter configured to perform non-adaptive hole blowing (usinga single, fixed low-magnitude threshold) on representative HSUPA signals(row labeled “HB w/o FIR”) to the ACLR5s, ACLR10s, R×N45 and EVM of apolar transmitter similar to that in FIG. 9 (except also including anon-linear phase filter in the PM path) configured to perform adaptivehole blowing (using a tri-level (α2, α2, α3) low-magnitude threshold αdefined by a three-step step function ƒ(S) similar to that shown in FIG.13) on the same HSUPA signals (row labeled “Adaptive HB w/o FIR”). TheACLR5, ACLR10, R×N45 and EVM performance for the same polar transmitter,but which also includes various combinations of the AM-path linear FIRfilter 3008 and a PM-path linear FIR filter 3010 are also included inthe table. The results show that applying adaptive hole blowing reducesACLR10 and R×N45 below that which is achievable without the benefit ofadaptive hole blowing, even without additional AM-path or PM-path linearfiltering, and without substantially increasing EVM. Further reductionin R×N with similar or better EVM performance is seen to be achieved byincluding either or both of the AM-path and PM-path linear FIR filters3008 and 3010.

The adaptive hole blowing methods and apparatus of the present inventionare well-suited for polar transmitters to reduce high frequency eventsin polar domain signals. They also compensate for PA memory effects andnonlinearity design characteristics. The adaptive hole blowing methodmaintains an “always on” transistor state to reduce PA memory effects ofentering and exiting “off” transistor states, and reduces PA dynamicrange usage at lowest output power where PA is highly nonlinear.

Though well-suited for polar transmitters, the adaptive hole blowingmethods and apparatus of the present invention may also be used in othertypes of transmitters, such as the quadrature-modulator-basedtransmitter 3200 shown in FIG. 32. The quadrature-modulator-basedtransmitter 3200 comprises a baseband modulator 3202, an AMR reductioncircuit 3204 (similar to the AMR reduction circuit 904 of the polartransmitter 900 in FIG. 9, except without the CORDIC converter 920) anda quadrature modulator 3206. The baseband modulator 3202 and AMRreduction circuit 3204 operate similar to how the baseband modulator 902and AMR reduction circuit 904 of the polar transmitter 900 operate (withthe exception of not converting the modified rectangular-coordinatebaseband signal ŝ(t)=Î(t)+j{circumflex over (Q)}(t) to polarcoordinates), so a similar description is not provided here.

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample, and not limitation. It will be apparent to persons skilled inthe relevant art that various changes in form and detail may be madetherein without departing from the spirit and scope of the invention.The scope of the invention should, therefore, be determined not withreference to the above description, but should instead be determinedwith reference to the appended claims, along with the full scope ofequivalents to which such claims are entitled.

1. A communications transmitter, comprising: a symbol mapper configuredto generate a sequence of symbols from a binary-source data streamcontaining a message to be transmitted; a pulse-shaping filterconfigured to generate a baseband signal based on said sequence ofsymbols; an average-to-minimum magnitude ratio (AMR) reduction circuitconfigured to compare magnitudes of samples of said baseband signal to atime-varying low-magnitude threshold that is varied at a rate equal toor dependent on a rate at which the symbol mapper generates the sequenceof symbols, and modify said baseband signal to produce a modifiedbaseband signal having a reduced AMR if a sample of said baseband signalis determined to have a magnitude less than said time-varyinglow-magnitude threshold; and a modulator configured to modulate acarrier signal based on modulation information contained in saidmodified baseband signal.
 2. A method of reducing the average-to-minimummagnitude ratio (AMR) of a baseband signal in a communicationstransmitter, comprising: generating a sequence of symbols from abinary-source data stream containing a message to be transmitted; pulseshaping said sequence of symbols to form a baseband signal; samplingsaid baseband signal; setting a time-varying low-magnitude threshold;varying the time-varying low-magnitude threshold at a rate equal to ordependent on a rate at which the sequence of symbols are generated;detecting whether samples of said baseband signal have a magnitude lessthan said time-varying low-magnitude threshold; and upon detecting asample of said baseband signal having a magnitude less than saidtime-varying threshold, modifying said baseband signal to produce amodified baseband signal having a lower AMR than said baseband signal.3. A baseband circuit for a communications transmitter, comprising: athreshold setting circuit configured to generate a time-varyinglow-magnitude threshold at a rate equal to or dependent on a rate atwhich symbols representing the baseband signal are generated or receivedby the threshold setting circuit; a local minimum detector configured tocompare magnitudes of samples of a baseband signal to said time-varyinglow-magnitude threshold; a circuit configured to generate an insertionpulse in response to said local minimum detector detecting alow-magnitude sample having a magnitude less than said time-varyinglow-magnitude threshold, and combine said insertion pulse with saidbaseband signal in the temporal vicinity of the detected low-magnitudesample to produce a modified baseband signal having a loweraverage-to-minimum magnitude ratio than that of said baseband signal. 4.The communications transmitter of claim 1 wherein said AMR reductioncircuit includes a threshold setting circuit configured to generate saidtime-varying low-magnitude threshold as a function of one or moresymbols in said sequence of symbols.
 5. The communications transmitterof claim 4 wherein said function depends on a magnitude or magnitudes ofsaid one or more symbols.
 6. The communications transmitter of claim 4wherein said function is a linear function of said one or more symbols.7. The communications transmitter of claim 4 wherein said function is anon-linear function of said one or more symbols.
 8. The communicationstransmitter of claim 1 wherein said AMR reduction circuit includes athreshold setting circuit configured to generate said time-varyinglow-magnitude threshold as a function of one or more samples of saidbaseband signal.
 9. The communications transmitter of claim 8 whereinsaid function depends on a magnitude or magnitudes of said one or moresamples.
 10. The communications transmitter of claim 8 wherein saidfunction is a linear function of said one or more samples.
 11. Thecommunications transmitter of claim 8 wherein said function is anon-linear function of said one or more samples.
 12. The communicationstransmitter of claim 1 wherein said modulator comprises a polarmodulator.
 13. The communications transmitter of claim 1 wherein saidmodulator comprises a quadrature modulator.
 14. The method of claim 2wherein said time-varying low-magnitude threshold is a function of oneor more symbols in said sequence of symbols.
 15. The method of claim 14wherein said function depends on a magnitude or magnitudes of said oneor more symbols.
 16. The method of claim 14 wherein said function is alinear function of said one or more symbols.
 17. The method of claim 14wherein said function is a non-linear function of said one or moresymbols.
 18. The method of claim 14 wherein said time-varyinglow-magnitude threshold is a function of one or more samples of saidbaseband signal.
 19. The method of claim 18 wherein said functiondepends on a magnitude or magnitudes of said one or more samples. 20.The method of claim 18 wherein said function is a linear function ofsaid one or more samples.
 21. The method of claim 18 wherein saidfunction is a non-linear function of said one or more samples.
 22. Themethod of claim 2, further comprising varying said time-varyinglow-magnitude threshold at a rate equal to or dependent on a rate atwhich said sequence of symbols are generated.
 23. The method of claim 2,further comprising varying said time-varying low-magnitude threshold ata rate equal to or dependent on a rate at which said baseband signal issampled.
 24. The method of claim 2, further comprising upconverting saidmodified baseband signal using a quadrature modulator.
 25. The basebandcircuit of claim 3 wherein said threshold setting circuit is configuredto receive symbols representing said baseband signal and vary saidtime-varying low-magnitude threshold according to a function thatdepends on one or more symbols of said symbols.
 26. The baseband circuitof claim 25 wherein said function depends on a magnitude or magnitudesof said one or more symbols.
 27. The baseband circuit of claim 25wherein said function is a linear function of said one or more symbols.28. The baseband circuit of claim 25 wherein said function is anon-linear function of said one or more symbols.
 29. The basebandcircuit of claim 3 wherein said threshold setting circuit is configuredto vary said time-varying low-magnitude threshold according to afunction that depends on one or more samples of said baseband signal.30. The baseband circuit of claim 29 wherein said function depends on amagnitude or magnitudes of said one or more samples.
 31. The basebandcircuit of claim 29 wherein said function is a linear function of saidone or more samples.
 32. The baseband circuit of claim 29 wherein saidfunction is a non-linear function of said one or more samples.